1. Field of the Invention
The present invention relates to a switching power supply circuit.
2. Description of the Related Art
As types of a so-called soft switching power supply that employs a resonant converter, a current resonant type and a voltage resonant type have been widely known. Currently, half-bridge current resonant converters formed of a two-transistor switching element have been widely employed since they can easily be put into practical use.
However, since characteristics of high-breakdown-voltage switching elements are currently being improved for example, problems about breakdown voltage associated with putting voltage resonant converters into practical use are being cleared up. Furthermore, it is known that a single-ended voltage resonant converter formed of a one-transistor switching element is advantageous over a one-transistor current resonant forward converter in terms of input feedback noise and noise components of a DC output voltage line.
FIG. 16 illustrates one configuration example of a switching power supply circuit including a single-ended voltage resonant converter. This voltage resonant converter is combined with a series resonant circuit on the secondary side to be described later, formed of a leakage inductor L2 of a secondary winding and a secondary-side series resonant capacitor C2, and hence is referred to as a multiple resonant converter.
In the switching power supply circuit of FIG. 16, a voltage from a commercial alternating-current power supply AC is rectified and smoothed by a rectifying and smoothing circuit formed of a bridge rectifier circuit Di and a smoothing capacitor Ci, to thereby produce a DC input voltage Ei as the voltage across the smoothing capacitor Ci. The lines from the commercial power supply AC are provided with a noise filter that includes a pair of common mode choke coils CMC and two across-line capacitors CL, and removes common mode noise.
The DC input voltage Ei is input to the voltage resonant converter as a DC input voltage. The voltage resonant converter has a single-ended configuration including a one-transistor switching element Q1 as described above. The voltage resonant converter in this circuit is separately excited. Specifically, the switching element Q1 formed of a MOSFET is switch-driven by an oscillation and drive circuit 2.
A body diode DD1 of the MOSFET is connected in parallel to the switching element Q1. In addition, a primary-side parallel resonant capacitor Cr is connected in parallel to the channel between the drain and source of the switching element Q1. The primary-side parallel resonant capacitor Cr and a leakage inductor L1 of a primary winding N1 in an isolation converter transformer PIT form a primary-side parallel resonant circuit (voltage resonant circuit). This primary-side parallel resonant circuit offers voltage resonant operation as the switching operation of the switching element Q1.
In order to switch-drive the switching element Q1, the oscillation and drive circuit 2 applies a gate voltage as a drive signal to the gate of the switching element Q1. Thus, the switching element Q1 implements switching operation with the switching frequency dependent upon the cycle of the drive signal.
The isolation converter transformer PIT transmits the switching output from the switching element Q1 to the secondary side. As shown in FIG. 17, the isolation converter transformer PIT is constructed of an EE-shaped core formed by combining E-shaped cores CR1 and CR2 composed of a ferrite material for example. Furthermore, the primary winding N1 and a secondary winding N2 are wound on a bobbin B that covers the center magnetic leg of the EE-shaped core, with the winding part being divided into the primary side and secondary side. In addition, a gap G with a length of about 0.8 to 1.0 mm is provided in the center leg of the EE-shaped core in the isolation converter transformer PIT, so that a coupling coefficient k of about 0.80 to 0.85 is obtained between the primary and secondary sides. When the coupling coefficient k has such a value, the coupling degree between the primary and secondary sides may be regarded as loose coupling, and thus it is correspondingly difficult to obtain the saturation state. The value of the coupling coefficient k is a factor in defining the leakage inductance (inductance of the leakage inductor L1).
The primary winding N1 in the isolation converter transformer PIT is interposed between the switching element Q1 and the positive electrode of the smoothing capacitor Ci, which allows the transmission of the switching output from the switching element Q1. In the secondary winding N2 of the isolation converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated.
On the secondary side, the secondary-side series resonant capacitor C2 is connected in series to one end of the secondary winding N2, and therefore the leakage inductor L2 of the secondary winding N2 and the capacitance of the secondary-side series resonant capacitor C2 form a secondary-side series resonant circuit (current resonant circuit).
Furthermore, rectifier diodes Do1 and Do2 and a smoothing capacitor Co are connected to this secondary-side series resonant circuit as shown in the drawing, so that a voltage-doubler half-wave rectifier circuit is formed. This voltage-doubler half-wave rectifier circuit produces, as the voltage across the smoothing capacitor Co, a DC output voltage Eo with the level twice that of a secondary winding voltage V3 induced in the secondary winding N2. The DC output voltage Eo is supplied to a load, and is input to a control circuit 1 as a detected voltage for constant-voltage control.
The control circuit 1 detects the level of the DC output voltage Eo input as a detected voltage, and then inputs the obtained detection output to the oscillation and drive circuit 2. The oscillation and drive circuit 2 outputs a drive signal of which frequency and so on are varied depending on the level of the DC output voltage Eo indicated by the detection output, to thereby control the switching operation of the switching element Q1 so that the DC output voltage Eo is kept constant at a predetermined level. Thus, stabilization control of the DC output voltage Eo is achieved.
FIGS. 18A to 18C and 19 show experimental results on the power supply circuit with the configuration shown in FIG. 16. For the experiments, major parts in the power supply circuit of FIG. 16 were designed to have the following parameters.
EER-35 was chosen as the core of the isolation converter transformer PIT, and a gap in the center leg thereof was designed to have a gap length of 1 mm. The numbers of turns of the primary winding N1 and the secondary winding N2 were set to 39 T and 23 T, respectively. The induction voltage level per one turn (T) in the secondary winding N2 was set to 3 V/T. The coupling coefficient k of the isolation converter transformer PIT was set to 0.81.
The capacitance of the primary-side parallel resonant capacitor Cr was set to 3900 pF (picofarad). The capacitance of the secondary-side series resonant capacitor C2 was set to 0.1 μF (microfarad). Accordingly, the primary-side parallel resonant frequency fo1 of the primary-side parallel resonant circuit was set to 230 kHz (kilohertz), and the secondary-side series resonant frequency fo2 of the secondary-side series resonant circuit was set to 82 kHz. Therefore, the relative relationship between the primary-side parallel resonant frequency fo1 and the secondary-side series resonant frequency fo2 can be represented as fo1≈2.8×fo2.
The rated level of the DC output voltage Eo was 135 V. The allowable load power range was from the maximum load power Pomax of 200 W to the minimum load power Pomin of 0 W.
FIGS. 18A to 18C are waveform diagrams showing the operation of the major parts in the power supply circuit shown in FIG. 16, with reflecting the corresponding switching cycle of the switching element Q1. FIG. 18A shows a switching voltage V1 applied to the switching element Q1, a switching current IQ1, a primary winding current I2, a secondary winding current I3, and rectified currents ID1 and ID2, when the load power is the maximum load power Pomax of 200 W. FIG. 18B shows the switching voltage V1, the switching current IQ1, the primary winding current I2, and the secondary winding current I3, when the load power is intermediate load power Po of 120 W. FIG. 18C shows the switching voltage V1 and the switching current IQ1 when the load power is the minimum load power Pomin of 0 W.
The switching voltage V1 is the voltage obtained across the switching element Q1, and has a waveform like ones illustrated in FIGS. 18A to 18C. Specifically, the voltage level is at the zero level during periods TON that are the periods when the switching element Q1 is in the ON-state, while a sinusoidal resonant pulse is obtained during periods TOFF that are the periods when it is in the OFF-state. This resonant pulse waveform of the switching voltage V1 indicates that the operation of the primary-side switching converter is voltage resonant operation.
The switching current IQ1 is the current flowing through the switching element Q1 (and the body diode DD1). The switching current IQ1 flows with the illustrated waveform during the period TON, while it is at the zero level during the period TOFF.
The primary winding current I2 flowing through the primary winding N1 is the current resulting from the synthesis between the current flowing as the switching current IQ1 during the period TON and the current flowing to the primary-side parallel resonant capacitor Cr during the period TOFF. The rectified currents ID1 and ID2, shown only in FIG. 18A, flowing through the rectifier diodes Do1 and Do2 as the operation of the secondary-side rectifier circuit have sinusoidal waveforms like the illustrated ones. In the waveform diagrams, the waveform of the rectified current ID1 indicates the resonant operation of the secondary-side series resonant circuit more dominantly than the waveform of the rectified current ID2.
The secondary winding current I3 flowing through the secondary winding N2 has a waveform resulting from the synthesis between the waveforms of the rectified currents ID1 and ID2. FIG. 19 shows, as functions of the load, the switching frequency fs, the lengths of the periods TON and TOFF of the switching element Q1, and the AC to DC power conversion efficiency (ηAC→DC), regarding the power supply circuit shown in FIG. 16.
Referring initially to the AC to DC power conversion efficiency (ηAC→DC), it is apparent that high efficiency of 90% or more is achieved in a wide range of the load power Po from 50 W to 200 W. The inventor of the present application has previously confirmed, based on experiments, that such a characteristic is obtained when a single-ended voltage resonant converter is combined with a secondary-side series resonant circuit.
In addition, the switching frequency fs, the period TON, and the period TOFF in FIG. 19 indicate the switching operation of the power supply circuit in FIG. 16 as the characteristic of constant-voltage control against load variation. In this power supply circuit, the switching frequency fs is almost constant against the load variation. In contrast, the periods TON and TOFF show linear changes with opposite tendencies as shown in FIG. 19. This indicates that control of the switching operation against the variation of the DC output voltage Eo is implemented in such a manner that the time ratio between the ON and OFF periods is changed with the switching frequency (switching cycle) being kept almost constant. Such control can be regarded as pulse width modulation (PWM) control, in which the lengths of the ON and OFF periods within one switching cycle are changed. That is, the power supply circuit in FIG. 16 uses the PWM control for stabilization of the DC output voltage Eo.
FIG. 20 schematically shows the constant-voltage control characteristic of the power supply circuit shown in FIG. 16, based on the relationship between the switching frequency fs (kHz) and the DC output voltage Eo.
The power supply circuit shown in FIG. 16 includes a primary-side parallel resonant circuit and a secondary-side series resonant circuit, and therefore has two resonant impedance characteristics in a complex manner: the resonant impedance characteristic corresponding to the primary-side parallel resonant frequency fo1 of the primary-side parallel resonant circuit, and that corresponding to the secondary-side series resonant frequency fo2 of the secondary-side series resonant circuit. Since the power supply circuit in FIG. 16 has the frequency relationship fo1≈2.8×fo2, the secondary-side series resonant frequency fo2 is lower than the primary-side parallel resonant frequency fo1 also as shown in FIG. 20.
The characteristic curves in FIG. 20 show a constant-voltage control characteristic that depends on control of the switching frequency fs and is assumed based on these resonant frequencies and the condition of a certain constant input AC voltage VAC. Specifically, Characteristic curves A and B correspond to the maximum load power Pomax and the minimum load power Pomin, respectively, and indicate the constant-voltage control characteristic in relation to the resonant impedance corresponding to the primary-side parallel resonant frequency fo1 of the primary-side parallel resonant circuit. Characteristic curves C and D correspond to the maximum load power Pomax and the minimum load power Pomin, respectively, and indicate the constant-voltage control characteristic in relation to the resonant impedance corresponding to the secondary-side series resonant frequency fo2 of the secondary-side series resonant circuit. When, under the characteristic shown in FIG. 20, constant-voltage control is intended so that the output voltage is kept at the voltage tg that is the rated level of the DC output voltage Eo, the variation range of the switching frequency fs necessary for the constant-voltage control (requisite control range) can be expressed by the section indicated as Δfs.
The control range Δfs shown in FIG. 20 is from the frequency offering the voltage level tg on Characteristic curve C, corresponding to the secondary-side series resonant frequency fo2 of the secondary-side series resonant circuit and the maximum load power Pomax, to that on Characteristic curve B, corresponding to the primary-side parallel resonant frequency fo1 of the primary-side parallel resonant circuit and the minimum load power Pomin. The range Δfs intersects with Characteristic curve D, which corresponds to the secondary-side series resonant frequency fo2 of the secondary-side series resonant circuit and the minimum load power Pomin, and with Characteristic curve A, which corresponds to the primary-side parallel resonant frequency fo1 of the primary-side parallel resonant circuit and the maximum load power Pomax.
Therefore, as constant-voltage control operation, the power supply circuit in FIG. 16 implements switching drive control based on PWM control in which the time ratio in one switching cycle (ratio between the periods TON and TOFF) is changed with the switching frequency fs being kept almost constant. The implementation of the PWM control is indicated also by FIGS. 18A to 18C, in which the widths of the periods TOFF and TON change depending on the load power while the length of one switching cycle (TOFF+TON) at a time of Pomax=200 W and Po=120 W is almost constant irrespective of the load power.
This operation is due to such a resonant impedance characteristic of the power supply circuit against load variation that transition is implemented, in the narrow switching frequency range Δfs, between the state where the resonant impedance corresponding to the primary-side parallel resonant frequency fo1 of the primary-side parallel resonant circuit (capacitive impedance) is dominant, and the state where the resonant impedance corresponding to the secondary-side series resonant frequency fo2 of the secondary-side series resonant circuit (inductive impedance) is dominant.
The related art of the invention is disclosed in e.g. Japanese Patent Laid-open No. 2000-134925.